Processing for improved performance and reduced pilot

ABSTRACT

A digital spread spectrum communication system employing pilot-aided coherent multipath demodulation effects a substantial reduction in global-pilot and assigned-pilot overheads. The system and method uses a QPSK-modulated data signal whereby the modulated data is removed and the recovered carrier is used for channel amplitude and phase estimation. The resulting signal has no data modulation and is used as a pseudo-pilot signal. In conjunction with the pseudo-pilot signal, a multiple-input phase-locked loop is employed further eliminating errors due to carrier-offset by using a plurality of pseudo-pilot signals. A pilot signal is required to resolve absolute phase ambiguity, but at a greatly reduced magnitude.

CROSS REFERENCE TO RELATED APPLICATIONS

[0001] This application is a continuation of application Ser. No.09/078,417, filed on May 14, 1998.

BACKGROUND

[0002] 1. Field of the Invention

[0003] The present invention relates generally to digitalcommunications. More specifically, the invention relates to a system forand method of using a code division multiple access air interface whichgreatly reduces the signal power required for the global andassigned-pilots while improving performance by using the quadraturephase shift keyed (QPSK) traffic signal for a particular channel toperform channel estimation and carrier recovery.

[0004] 2. Description of the Prior Art

[0005] Most advanced communication technology today makes use of digitalspread spectrum modulation or code divisional multiple access (CDMA).Digital spread spectrum is a communication technique in which data istransmitted with a broadened band (spread spectrum) by modulating thedata to be transmitted with a pseudo-noise signal. CDMA can transmitdata without being affected by signal distortion or an interferingfrequency in the transmission path.

[0006] Shown in FIG. 1 is a simplified CDMA communication system thatinvolves a single communication channel of a given bandwidth which ismixed by a spreading code which repeats a predetermined patterngenerated by a pseudo-noise (pn) sequence generator. A data signal ismodulated with the pn sequence producing a digital spread spectrumsignal. A carrier signal is then modulated with the digital spreadspectrum signal establishing a forward link, and transmitted. A receiverdemodulates the transmission extracting the digital spread spectrumsignal. The transmitted data is reproduced after correlation with thematching pn sequence. The same process is repeated to establish areverse link.

[0007] During terrestrial communication, a transmitted signal isdisturbed by reflection due to varying terrain and environmentalconditions and man-made obstructions. This produces a plurality ofreceived signals with differing time delays at the receiver. This effectis commonly known as multipath propagation. Moreover, each path arrivesdelayed at the receiver with a unique amplitude and carrier phase.

[0008] To identify the multiple components in the multipath propagation,the relative delays and amplitudes and phases must be determined. Thisdetermination can be performed with a modulated data signal, buttypically, a more precise rendering is obtained when compared to anunmodulated signal. In most digital spread spectrum systems, it is moreeffective to use an unmodulated pilot signal discrete from thetransmitted modulated data by assigning the pilot an individual pnsequence. A global-pilot signal is most valuable on systems where manysignals are transmitted from a base station to multiple users.

[0009] In the case of a base station which is transmitting manychannels, the global-pilot signal provides the same pilot sequence tothe plurality of users serviced by that particular base station and isused for the initial acquisition of an individual user and for the userto obtain channel-estimates for coherent reception and for the combiningof the multipath components. However, at the required signal strength,the global-pilot signal may use up to 10 percent of the forwarddirection air capacity.

[0010] Similar multipath distortion affects a user's reverse linktransmission to the base station. Inserting in each individual user'sreturn signal an assigned-pilot may consume up to 20 percent of thetotal reverse channels air capacity.

[0011] Without phase and amplitude estimation, noncoherent ordifferentially coherent reception techniques must be performed.Accordingly, there exists a need for a coherent demodulation system thatreduces the air capacity of the global-pilot and assigned-pilot signalswhile maintaining the desired air-interface performance.

SUMMARY

[0012] The present invention relates to a digital spread spectrumcommunication system that employs pilot-aided coherent multipathdemodulation with a substantial reduction in global-pilot andassigned-pilot overheads. The system and method uses a QPSK- modulateddata signal whereby the modulated data is removed and the recoveredcarrier is used for channel amplitude and phase estimation. Theresulting signal has no data modulation and is used as a pseudo-pilotsignal. In conjunction with the pseudo-pilot signal, a multiple-inputphase-locked loop is employed further eliminating errors due tocarrier-offset by using a plurality of pseudo-pilot signals. A pilotsignal is still required to resolve absolute phase ambiguity, but at agreatly reduced magnitude.

[0013] Accordingly, it is an object of the present invention to providea code division multiple access communication system which reduces therequired global and assigned-pilot signal strength.

[0014] It is a further object of the invention to reduce the transmittedlevels of the global and assigned-pilots such that they consumenegligible overhead in the air interface while providing informationnecessary for coherent demodulation.

[0015] Other objects and advantages of the system and method will becomeapparent to those skilled in the art after reading the detaileddescription of the preferred embodiment.

BRIEF DESCRIPTION OF THE DRAWINGS

[0016]FIG. 1 is a simplified block diagram of a typical, prior art, CDMAcommunication system.

[0017]FIG. 2 is a detailed block diagram of a B-CDMA™ communicationsystem.

[0018]FIG. 3A is a plot of an in-phase bit stream.

[0019]FIG. 3B is a plot of a quadrature bit stream.

[0020]FIG. 3C is a plot of a pseudo-noise (pn) bit sequence.

[0021]FIG. 4 is a detailed block diagram of the present invention usingone pseudo-pilot signal, with carrier-offset correction implemented atthe chip level.

[0022]FIG. 5 is a block diagram of a rake receiver.

[0023]FIG. 6 is a diagram of a received symbol p_(o) on the QPSKconstellation showing a hard decision.

[0024]FIG. 7 is a diagram of the angle of correction corresponding tothe assigned symbol.

[0025]FIG. 8 is a diagram of the resultant symbol error after applyingthe correction corresponding to the assigned symbol.

[0026]FIG. 9 is a block diagram of a conventional phase-locked loop.

[0027]FIG. 10 is a detailed block diagram of the present invention usinga pseudo-pilot signal with carrier-offset correction implemented at thesymbol level.

[0028]FIG. 11 is a detailed block diagram of the present invention usinga pseudo-pilot signal and the MIPLL, with carrier-offset correctionimplemented at the chip level.

[0029]FIG. 12 is a block diagram of the multiple input phase-locked loop(MIPLL).

[0030]FIG. 13 is a detailed block diagram of the present invention usinga pseudo-pilot signal and the MIPLL, with carrier-offset correctionimplemented at the symbol level.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0031] The preferred embodiment will be described with reference to thedrawing figures where like numerals represent like elements throughout.

[0032] A B-CDMA™ communication system 25 as shown in FIG. 2 includes atransmitter 27 and a receiver 29, which may reside in either a basestation or a mobile user receiver. The transmitter 27 includes a signalprocessor 31 which encodes voice and nonvoice signals 33 into data atvarious rates, e.g. data rates of 8 kbps, 16 kbps, 32 kbps, or 64 kbps.The signal processor 31 selects a rate in dependence upon the type ofsignal, or in response to a set data rate.

[0033] By way of background, two steps are involved in the generation ofa transmitted signal in a multiple access environment. First, the inputdata 33 which can be considered a bi-phase modulated signal is encodedusing forward error-correcting coding (FEC) 35. For example, if a R=½convolution code is used, the single bi-phase modulated data signalbecomes bivariate or two bi-phase modulated signals. One signal isdesignated the in-phase channel I 41 a. The other signal is designatedthe quadrature channel Q 41 b. A complex number is in the form a+bj,where a and b are real numbers and j²=−1. Bi-phase modulated I and Qsignals are usually referred to as quadrature phase shift keying (QPSK).In the preferred embodiment, the tap generator polynomials for aconstraint length of K=7 and a convolutional code rate of R=½ areG₁=171₈37 and G₂=133₈39.

[0034] In the second step, the two bi-phase modulated data or symbols 41a, 41 b are spread with a complex pseudo-noise (pn) sequence. Theresulting I 45 a and Q 45 b spread signals are combined 53 with otherspread signals (channels) having different spreading codes, multiplied(mixed) with a carrier signal 51, and transmitted 55. The transmission55 may contain a plurality of individual channels having different datarates.

[0035] The receiver 29 includes a demodulator 57 a, 57 b which mixesdown the transmitted broadband signal 55 into an intermediate carrierfrequency 59 a, 59 b. A second down conversion reduces the signal tobaseband. The QPSK signal is then filtered 61 and mixed 63 a, 63 b withthe locally generated complex pn sequence 43 a, 43 b which matches theconjugate of the transmitted complex code. Only the original waveformswhich were spread by the same code at the transmitter 27 will beeffectively despread. Others will appear as noise to the receiver 29.The data 65 a, 65 b is then passed onto a signal processor 59 where FECdecoding is performed on the convolutionally encoded data.

[0036] As shown in FIGS. 3A and 3B, a QPSK symbol consists of one biteach from both the in-phase (I) and quadrature (Q) signals. The bits mayrepresent a quantized version of an analog sample or digital data. Itcan be seen that symbol duration t_(S) is equal to bit duration.

[0037] The transmitted symbols are spread by multiplying the QPSK symbolstream by a unique complex pn sequence. Both the I and Q pn sequencesare comprised of a bit stream generated at a much higher rate, typically100 to 200 times the symbol rate. One such pn sequence is shown in FIG.3C. The complex pn sequence is mixed with the complex-symbol bit streamproducing the digital spread signal. The components of the spread signalare known as chips having a much smaller duration t_(c).

[0038] When the signal is received and demodulated, the baseband signalis at the chip level. Both the I and Q components of the signal aredespread using the conjugate of the pn sequence used during spreading,returning the signal to the symbol level. However, due tocarrier-offset, phase corruption experienced during transmissionmanifests itself by distorting the individual chip waveforms. Ifcarrier-offset correction is performed at the chip level, it can be seenthat overall accuracy increases due to the inherent resolution of thechip-level signal. Carrier-offset correction may also be performed atthe symbol level, but with less overall accuracy. However, since thesymbol rate is much less than the chip rate, less overall processingspeed is required when the correction is done at the symbol level.

[0039] System architectures for receivers taught in accordance with thesystem and method of the present invention that do not require largemagnitude pilot signals follow. The following systems replace thefiltering, despreading and signal processing shown in FIG. 2. Thesystems are implemented with carrier-offset correction at both the chipand symbol levels.

[0040] As shown in FIG. 4, a receiver using the system 75 and method ofthe present invention is shown. A complex baseband digital spreadspectrum signal 77 comprised of in-phase and quadrature phase componentsis input and filtered using an adaptive matched filter (AMF) 79 or otheradaptive filtering means. The AMF 79 is a transversal filter (finiteimpulse response) which uses filter coefficients 81 to overlay delayedreplicas of the received signal 77 onto each other to provide a filteredsignal 83 having an increased signal-to-noise ratio (SNR). The output 83of the AMF 79 is coupled to a plurality of channel despreaders 85 ₁, 85₂, 85 _(n) and a pilot despreader 87. In the preferred embodiment, n=3.The pilot signal 89 is despread with a separate despreader 87 and pnsequence 91 contemporaneous with the transmitted data 77 assigned tochannels which are despread 85 ₁, 85 ₂, 85 _(n) with pn sequences 93 ₁,93 ₂, 93 _(n) of their own. After the data channels are despread 85 ₁,85 ₂, 85 _(n), the data bit streams 95 ₁, 95 ₂, 95 _(n) are coupled toViterbi decoders 97 ₁, 97 ₂, 97 _(n) and output 99 ₁, 99 ₂, 99 _(n).

[0041] The filter coefficients 81, or weights, used in adjusting the AMF79 are obtained by the demodulation of the individual multipathpropagation paths. This operation is performed by a rake receiver 101.The use of a rake receiver 101 to compensate for multipath distortion iswell known to those skilled in the communication arts.

[0042] As shown in FIG. 5, the rake receiver 101 consists of a parallelcombination of path demodulators (“fingers”) 103 ₀, 103 ₁, 103 ₂, 103_(n) which demodulate a particular multipath component. The pilotsequence tracking loop of a particular demodulator is initiated by thetiming estimation of a given path as determined by a pn sequence 105. Inthe prior art, a pilot signal is used for despreading the individualsignals of the rake. In this embodiment of the present invention, the pnsequence 105 may belong to any channel 93 ₁ of the communication system.The channel with the largest received signal is typically used.

[0043] Each path demodulator includes a complex mixer 107 ₀, 107 ₁, 107₂, 107 _(n), and summer and latch 109 ₀, 109 ₁, 109 ₂, 109 _(n). Foreach rake element, the pn sequence 105 is delayed τ 111 ₁, 111 ₂, 111_(n) by one chip and mixed 107 ₁, 107 ₂, 107 _(n) with the basebandspread spectrum signal 113 thereby despreading each signal. Eachmultiplication product is input into an accumulator 109 ₀, 109 ₁, 109 ₂,109 _(n) where it is added to the previous product and latched out afterthe next symbol-clock cycle. The rake receiver 101 provides relativepath values for each multipath component. The plurality of n-dimensionoutputs 115 ₀, 115 ₁, 115 ₂, 115 _(n) provide estimates of the sampledchannel impulse response that contain a relative phase error of either0°, 90°, 180°, or 270°.

[0044] Referring back to FIG. 4, the plurality of outputs from the rakereceiver are coupled to an n-dimensional complex mixer 117. Mixed witheach rake receiver 101 output 115 is a correction to remove the relativephase error contained in the rake output.

[0045] A pilot signal is also a complex QPSK signal, but with thequadrature component set at zero. The error correction 119 signal of thepresent invention is derived from the despread channel 95 ₁ by firstperforming a hard decision 121 on each of the symbols of the despreadsignal 95 ₁. A hard decision processor 121 determines the QPSKconstellation position that is closest to the despread symbol value.

[0046] As shown in FIG. 6, the Euclidean distance processor compares areceived symbol p_(o) of channel 1 to the four QPSK constellation pointsX_(1, 1), X_(−1, 1) X_(−1, −1), X_(1, −1). It is necessary to examineeach received symbol p_(o) due to corruption during transmission 55 bynoise and distortion, whether multipath or radio frequency. The harddecision processor 121 computes the four distances d₁, d₂, d₃, d₄ toeach quadrant from the received symbol p_(o) and chooses the shortestdistance d₂ and assigns that symbol location x_(−1, 1). The originalsymbol coordinates p_(o) are discarded.

[0047] Referring back to FIG. 4, after undergoing each hard symboldecision 121, the complex conjugates 123 for each symbol output 125 aredetermined. A complex conjugate is one of a pair of complex numbers withidentical real parts and with imaginary parts differing only in sign.

[0048] As shown in FIG. 7, a symbol is demodulated or derotated by firstdetermining the complex conjugate of the assigned symbol coordinatesx_(−1, −1), forming the correction signal 119 which is used to removethe relative phase error contained in the rake output. Thus, the rakeoutput is effectively derotated by the angle associated with the harddecision, removing the relative phase error. This operation effectivelyprovides a rake that is driven by a pilot signal, but without anabsolute phase reference.

[0049] Referring back to FIG. 4, the output 119 from the complexconjugate 123 is coupled to a complex n-dimensional mixer 117 where eachoutput of the rake receiver 101 is mixed with the correction signal 119.The resulting products 127 are noisy estimates of the channel impulseresponse P₁ as shown in FIG. 8. The error shown in FIG. 8 is indicatedby a radian distance of τ/6 from the in-phase axis.

[0050] Referring back to FIG. 4, the outputs 129 of the complexn-dimensional mixer 117 are coupled to an n-dimensional channelestimator 131. The channel estimator 131 is a plurality of low-passfilters filtering each multipath component. The outputs of then-dimensional mixer 117 are coupled to the AMF 79. These signals act asthe AMF 79 filter weights. The AMF 79 filters the baseband signal tocompensate for channel distortion due to multipath without requiring alarge magnitude pilot signal.

[0051] Rake receivers 101 are used in conjunction with phase-locked loop(PLL) 133 circuits to remove carrier-offset. Carrier-offset occurs as aresult of transmitter/receiver component mismatches and other RFdistortion. The present invention 75 requires that a low level pilotsignal 135 be produced by despreading 87 the pilot from the basebandsignal 77 with a pilot pn sequence 91. The pilot signal is coupled to asingle input PLL 133. The PLL 133 measures the phase difference betweenthe pilot signal 135 and a reference phase of 0. The despread pilotsignal 135 is the actual error signal coupled to the PLL 133.

[0052] A conventional PLL 133 is shown in FIG. 9. The PLL 133 includesan arctangent analyzer 136, complex filter 137, an integrator 139 and aphase-to-complex-number converter 141. The pilot signal 135 is the errorsignal input to the PLL 133 and is coupled to the complex filter 137.The complex filter 137 includes two gain stages, an integrator 145 and asummer 147. The output from the complex filter is coupled to theintegrator 139. The integral of frequency is phase, which is output 140to the converter 141. The phase output 140 is coupled to a converter 141which converts the phase signal into a complex signal for mixing 151with the baseband signal 77. Since the upstream operations arecommutative, the output 149 of the PLL 133 is also the feedback loopinto the system 75.

[0053] By implementing the hard decision 121 and derotation 123 of thedata modulation, the process provides channel estimation without the useof a large pilot signal. If an error occurs during the hard decisionprocess and the quadrant of the received data symbol is not assignedcorrectly, the process suffers a phase error. The probability of phaseerror is reduced, however, due to the increased signal-to-noise ratio ofthe traffic channel. The errors that occur are filtered out during thechannel-estimation and carrier-recovery processes. The traffic channelis approximately 6 dB stronger (2×) than the level of the despreadpilot.

[0054] As described earlier, the present invention can also be performedwith carrier-offset correction at the symbol level. An alternativeembodiment 150 implemented at the symbol level is shown in FIG. 10. Thedifference between the chip and symbol level processes occur where theoutput of the conventional PLL 133 is combined. At the symbol level, thePLL output 140 does not undergo chip conversion 141 and is introducedinto the AMF 79 weights after the rake receiver 101 by anothern-dimensional mixer 153. The phase correction 140 feedback must also bemixed 154 ₁, 154 ₂, 154 _(n) with the outputs 95 ₁, 95 ₂, 95 _(n) ofeach of the plurality of channel despreaders 85 ₁, 85 ₂, 85 _(n) andmixed 156 with the output 135 of the pilot despreader 87.

[0055] As shown in FIG. 11, another alternative embodiment 193 uses avariation of the earlier embodiments whereby a hard decision is renderedon each received symbol after despreading and derotated by a radianamount equal to the complex conjugate. The alternate approach 193 uses aplurality of channel despreaders 85 ₁, 85 ₂, 85 _(n) and the pilotdespreader 87 as inputs to a multiple input phase-locked loop (MIPLL)157 shown in FIG. 12. Since each of the despread channels 95 ₁, 95 ₂, 95_(n) contains an ambiguous representation of the pilot signal, a smallsignal pilot 135 is required to serve as an absolute reference. Thedespread symbols from all channels in conjunction with the despreadsmall signal pilot signal are input to the MIPLL 157.

[0056] Referring to FIG. 12, the output from each channel 95 ₁, 95 ₂, 95_(n) is coupled to a hard decision/complex conjugate operation 159 ₁,159 ₂, 159 _(n). The derotated pseudo-pilots 161 ₁, 161 ₂, 161 _(n) arethen mixed with the delayed symbols producing a complex voltage error163 ₁, 163 ₂, 163 _(n). The error 165 ₁, 165 ₂, 165 _(n) is input into aconverter 167 ₁, 167 ₂, 167 _(n), 167 _(n+1) which takes an inversetangent converting the complex number into a phase error 169 ₁, 169 ₂,169 _(n), 169 _(n+1). Each phase error 169 ₁, 169 ₂, 169 _(n), 169_(n+1) is input into a maximum likelihood combiner 171 which assignsvarious weights to the plurality of inputs and produces a sum output.Also included in the sum is the small signal pilot 135 phase 169 _(n+1)which is despread 135 and converted 167 _(n+1). The weighting of thesmall pilot signal may be emphasized since its phase is unambiguous.

[0057] The output of the combiner 173 is the estimate of thecarrier-offset and is coupled to a complex filter 175 and coupled to anintegrator 177. All channels contribute to the estimate of thecarrier-offset frequency with the absolute phase error removed by theunambiguous pilot signal. The integrator accumulates the history of thesummed signal over many samples. After integration, the estimate of thephase error is output 179 converted to a complex voltage and output 183.

[0058] Referring back to FIG. 11, the output 183 of the MIPLL 157 iscoupled to a complex mixer 185 upstream of the rake receiver. Thiscompletes the error feedback for the MIPLL 157. Even though thisembodiment requires additional resources and complexity, the MIPLL 157architecture can be efficiently implemented and executed in a digitalsignal processor (DSP).

[0059] Referring now to the alternative embodiment 195 shown in FIG. 13,this embodiment 195 mixes the output of the MIPLL 157 at the symbollevel. The MIPLL 157 is mixed 197 with the output of the rake receiver101. As described above, the output of the rake receiver 101 is at thesymbol level. The symbol-to-chip conversion 181 in the MIPLL 157architecture is disabled. Since the output 183 of the MIPLL 157 is mixedwith the outputs of the rake 101 which are used only for the AMF 79weights, the phase correction for carrier-offset must be added to theportion of the receiver that processes traffic data. A plurality ofmixers 199 ₁, 199 ₂, 199 _(n) downstream of each channel despreader 85₁, 85 ₂, 85 _(n) and a mixer 193 downstream of the pilot despreader 87are therefore required to mix the phase-corrected output 183 (at thesymbol level) as feedback into the system.

[0060] The present invention maintains the transmitted pilot signal at alow level to provide an absolute phase reference while reducing pilotinterference and increasing air capacity. The net effect is the virtualelimination of the pilot overhead.

[0061] While specific embodiments of the present invention have beenshown and described, many modifications and variations could be made byone skilled in the art without departing from the spirit and scope ofthe invention. The above description serves to illustrate and not limitthe particular form in any way.

What is claimed is:
 1. A receiver for use in a communication station ofa CDMA system wherein a plurality of communication stations communicatewith each other over a CDMA air interface using a plurality of channelsand a pilot signal for carrier-offset recovery during reception, thereceiver comprising: an adaptive matched filter for receivingdemodulated CDMA communication signals producing a filtered signal byusing a weighting signal; a rake receiver for receiving the demodulatedCDMA communication signals and a pseudo-noise signal generated for aselected channel and producing a filter weighting signal; means fordefining the filter weighting signal with a correction signal, saidcorrection signal to produce the weighting signal used by said adaptivematched filter; a channel despreader for said selected channel coupledto said adaptive matched filter output for despreading said filteredsignal using the pseudo-noise signal generated for said selected channelto produce a despread channel signal of said selected channel; a pilotchannel despreader for a pilot channel coupled to said adaptive matchedfilter output for despreading said filtered signal using a pseudo-noisesignal generator for said pilot channel to produce a despread pilotsignal of said pilot channel; a hard decision processor in associationwith a complex conjugate processor for receiving the despread channelsignal of said selected channel and producing said correction signal;and a phase-locked loop utilizing at least said despread pilot signalfor producing a phase correction signal which is applied to producephase-corrected channel signals.
 2. The receiver according to claim 1further comprising a plurality of channel despreaders, each coupled tosaid adaptive matched filter output for despreading said filtered signaleach using an associated pseudo-noise signal generator to produce aplurality of despread channel signals.
 3. The receiver according toclaim 2 wherein the number of channel despreaders is three.
 4. Thereceiver according to claim 2 wherein said phase-locked loop phasecorrection signal is at a chip level and is applied to said demodulatedCDMA communication signals.
 5. The receiver according to claim 2 whereineach of the plurality of channels is a complex, bi-phase modulatedsignal comprised of symbols including in-phase and quadrature componentsrepresenting data, said hard decision processor compares each despreadchannel signal symbol to one of four possible quadrature constellationpoints and assigns each of said symbols to a nearest constellationpoint, and said complex conjugate processor derotates each of saidsymbols by determining a complex conjugate of each of said assignedpoints to produce said correction signal.
 6. The receiver according toclaim 2 wherein said phase-locked loop further comprises a plurality ofinputs corresponding with said plurality of channel despreaders.
 7. Thereceiver according to claim 6 wherein said phase-locked loop furthercomprises: a hard decision processor in association with said complexconjugate processor with a local feedback loop for each of saidcorresponding channel despreader inputs to produce an error estimatesignal for a respective channel signal; each said error estimate signaland said despreader pilot signal coupled to an inverse tangent processorto produce a corresponding phase correction signal; and said respectivechannel phase correction signal and pilot phase correction signalcoupled to a maximum likelihood combiner producing a combinationcorrection signal coupled to an integrator to produce said phasecorrection signal.
 8. The receiver according to claim 7 wherein thenumber of channel despreaders is three.
 9. The receiver according toclaim 1 wherein said phase-locked loop phase correction signal is at asymbol level and is applied to said filter weighting signal and to saiddespread channel signals of said channel and pilot channel despreaders.10. The receiver according to claim 9 further comprising a plurality ofchannel despreaders, each coupled to said adaptive matched filter outputfor despreading said filtered signal using an associated pseudo-noisesignal generator to produce a plurality of despread channel signals. 11.The receiver according to claim 10 wherein the number of channeldespreaders is three.
 12. The receiver according to claim 10 whereinsaid phase-locked loop further comprises a plurality of signal inputscorresponding with said plurality of channel despreaders.
 13. Thereceiver according to claim 12 wherein said phase-locked loop furthercomprises: a hard decision processor in association with a complexconjugate processor with a local feedback loop for each of saidplurality of signal inputs, each producing an error estimate for arespective channel signal; each of said channel error estimates and saiddespreader pilot signal coupled to an inverse tangent processoroutputting a channel phase correction signal; and said channel and pilotphase correction signals coupled to a maximum likelihood combinerproducing a combination correction signal coupled to an integrator toproduce said phase correction signal.
 14. The receiver according toclaim 13 wherein the number of channel despreaders is three.
 15. Amethod of receiving at least one of a plurality of channels over a CDMAair interface using a reduced magnitude pilot signal for carrier-offsetrecovery during reception wherein a plurality of communication stationscommunicate with each other comprising the steps: receiving demodulatedCDMA communication signals; filtering said received demodulated CDMAcommunication signals with an adaptive matched filter to produce afiltered signal by using a weighting signal; producing a filterweighting signal with a rake receiver using said demodulated CDMAcommunication signals and a pseudo-noise signal generated for a selectedchannel; refining said filter weighting signal with a correction signal;despreading said selected channel from said filtered signal using thepseudo-noise signal for said selected channel to produce a despreadchannel signal of said selected channel; despreading a pilot channelfrom said filtered signal using a pseudo-noise signal generated for saidpilot channel to produce a despread pilot signal of said pilot channel;processing said selected despread channel signal with a hard decisionprocessor in association with a complex conjugate processor to producesaid correction signal; and generating a phase correction signal fromsaid despread pilot signal with a phased-locked loop to phase-correctsaid selected channel signal.
 16. The method according to claim 15wherein said phase correction signal is at a chip level.
 17. The methodaccording to claim 16 wherein the step of despreading said selectedchannel also includes despreading a plurality of channels to producedespread channel signals.
 18. The method according to claim 17 whereinsaid step of generating a phase correction signal further includes thesteps of: assigning a received symbol to one of four possible quadratureconstellation points for said despread selected channel signal and eachof said despread channel signals; derotating each of said assignedsymbols for said despread selected channel signal and each of saiddespread channel signals by determining the complex conjugate of each ofsaid assigned points to produce respective error estimate signals;coupling each of said error estimate signals and said despread pilotsignal to inverse tangent processors to produce corresponding phasecorrection signals; and combining said channel and pilot phasecorrection signals to produce said phase correction signal.
 19. Themethod according to claim 15 wherein said phase correction signal is ata symbol level.
 20. The method according to claim 19 wherein the step ofdespreading said selected channel also includes despreading a pluralityof channels to produce despread channel signals.
 21. The methodaccording to claim 20 wherein said step of generating a phase correctionsignal further includes the steps of: assigning a received symbol to oneof four possible quadrature constellation points for said despreadselected channel signal and each of said despread channel signals;derotating each of said assigned symbols for said despread selectedchannel signal and each of said despread channel signals by determiningthe complex conjugate of each of said assigned points to producerespective error estimate signals; coupling of each said error estimatesignals and said despread pilot signal to inverse tangent processors toproduce corresponding phase correction signals; and combining saidchannel and pilot phase correction signals to produce said phasecorrection signal.